Table 51. Design Parameters

10.2.1.2.1 Select R_SNS and CL Setting

LM5066I can be used with a VCL of 26 or 50 mV. Using the 26-mV threshold results in a lower R_SNS and lower I²R losses. The 26-mV option is selected for this design by connecting the CL pin directly to V_DD. TI recommends to target a current limit that is at least 10% above the maximum load current to account for the tolerance of the LM5066I current limit. Targeting a current limit of 11 A, the sense resistor can be computed as follows:

Equation 7.

Typically, sense resistors are only available in discrete values. If a precise current limit is desired, a sense resistor along with a resistor divider can be used as shown in Figure 20.

Figure 20. SENSE Resistor Divider

The next larger available sense resistor should be chosen (3 mΩ in this case). The ratio of R₁ and R₂ can be computed as follows:

Equation 8.

Note that the SENSE pin pulls 25 μA of current, which creates an offset across R₂. TI recommends to keep R₂ below 10 Ω to reduce the offset that this introduces. In addition, the 1% resistors add to the current monitoring error. Finally, if the resistor divider approach is used, the user should compute the effective sense resistance (R_SNS,EFF) using Equation 9 and use that in all equations instead of R_SNS.

Equation 9.

Note that for many applications, a precise current limit may not be required. In that case, it is simpler to pick the next smaller available sense resistor. For this application, a 2-mΩ resistor can be used for a 13-A current limit.

10.2.1.2.2 Selecting the Hotswap FETs

It is critical to select the correct MOSFET for a hotswap design. The device must meet the following requirements:

• The V_DS rating should be sufficient to handle the maximum system voltage along with any ringing caused by transients. For most 48-V systems, a 100-V FET is a good choice.
• The SOA of the FET should be sufficient to handle all usage cases: start-up, hot-short, and start into short.
• R_DSON should be sufficiently low to maintain the junction and case temperature below the maximum rating of the FET. In fact, TI recommends to keep the steady-state FET temperature below 125°C to allow margin to handle transients.
• Maximum continuous current rating should be above the maximum load current and the pulsed-drain current must be greater than the current threshold of the circuit breaker. Most MOSFETs that pass the first three requirements also pass these two.
• A V_GS rating of ±20 V is required because the LM5066I can pull up the gate as high as 16 V above source.

For this design, the PSMN4R8-100BSE was selected for its low R_DSON and superior SOA. After selecting the MOSFET, the maximum steady-state case temperature can be computed as follows:

Equation 10.

Note that the R_DSON is a strong function of junction temperature, which for most D2PACK MOSFETs is very close to the case temperature. A few iterations of the previous equations may be necessary to converge on the final R_DSON and T_C,MAX value. According to the PSMN4R8-100BSE data sheet, it's R_DSON doubles at 110°C. Equation 11 uses this R_DSON value to compute the T_C,MAX. Note that the computed T_C,MAX is close to the junction temperature assumed for R_DSON. Thus, no further iterations are necessary.

Equation 11.

10.2.1.2.3 Select Power Limit

In general, a lower power limit setting is preferred to reduce the stress on the MOSFET. However, when the LM5066I is set to a very-low power limit setting, it has to regulate the FET current and hence the voltage across the sense resistor (V_SNS) to a very-low value. V_SNS can be computed as shown in Equation 12.

Equation 12.

To avoid significant degradation of the power limiting, TI does not recommend a V_SNS of less than 4 mV. Based on this requirement, the minimum allowed power limit can be computed as follows:

Equation 13.

In most applications, the power limit can be set to P_LIM,MIN using Equation 14. Here R_SNS and R_PWR are in ohms and P_LIM is in watts.

Equation 14.

The closest available resistor should be selected. In this case, a 28.2-kΩ resistor was chosen.

10.2.1.2.4 Set Fault Timer

The fault timer runs when the hotswap is in power limit or current limit, which is the case during start-up. Thus, the timer has to be sized large enough to prevent a time-out during start-up. If the part starts directly into current limit (I_LIM × V_DS < P_LIM), the maximum start time can be computed with Equation 15.

Equation 15.

For most designs (including this example), I_LIM × V_DS > P_LIM, so the hotswap starts in power limit and transitions into current limit. In that case, the maximum start time can be computed as in Equation 16.

Equation 16.

Note that the above start-time is based on typical current limit and power limit values. To ensure that the timer never times out during start-up, TI recommends to set the fault time (t_flt) to be 1.5 × t_start,max or 5.1 ms. This accounts for the variation in power limit, timer current, and timer capacitance. Thus, C_TIMER can be computed as follows:

Equation 17.

The next largest standards capacitor value for C_TIMER is chosen as 100 nF. After C_TIMER is chosen, the actual programmed fault time can be computed as follows:

Equation 18.

10.2.1.2.5 Check MOSFET SOA

When the power limit and fault timer are chosen, it is critical to check that the FET stays within its SOA during all test conditions. During a hot-short the circuit breaker trips and the LM5066I restarts into power limit until the timer runs out. In the worst case, the MOSFET’s V_DS equals V_IN,MAX, I_DS equals P_LIM / V_IN,MAX and the stress event lasts for t_flt. For this design example, the MOSFET has 60 V, 2 A across it for 5.2 ms.

Based on the SOA of the PSMN4R8-100BSE, it can handle 60 V, 30 A for 1 ms and it can handle 60 V, 6 A for 10 ms. For 5.2 ms, the SOA can be extrapolated by approximating SOA versus time as a power function as shown below:

Equation 19.

Note that the SOA of a MOSFET is specified at a case temperature of 25°C, while the case temperature can be much hotter during a hot-short. The SOA should be de-rated based on T_C,MAX using Equation 20:

Equation 20.

Based on this calculation, the MOSFET can handle 3.85 A, 60 V for 5.2 ms at elevated case temperature, but is only required to handle 2 A during a hot-short. Thus, there is good margin and the design is robust. In general, TI recommends that the MOSFET can handle 1.3× more than what is required during a hot-short. This provides margin to account for the variance of the power limit and fault time.

10.2.1.2.6 Set UVLO and OVLO Thresholds

By programming the UVLO and OVLO thresholds, the LM5066I enables the series-pass device (Q₁) when the input supply voltage (V_IN) is within the desired operational range. If V_IN is below the UVLO threshold or above the OVLO threshold, Q₁ is switched off, denying power to the load. Hysteresis is provided for each threshold.

10.2.1.2.6.1 Option A

The configuration shown in Figure 21 requires three resistors (R1 to R3) to set the thresholds.

Figure 21. UVLO And OVLO Thresholds Set By R1-R3

The procedure to calculate the resistor values is as follows:

• Choose the upper UVLO threshold (V_UVH) and the lower UVLO threshold (V_UVL).
• Choose the upper OVLO threshold (V_OVH).
• The lower OVLO threshold (V_OVL) cannot be chosen in advance in this case, but is determined after the values for R1 to R3 are determined. If V_OVL must be accurately defined in addition to the other three thresholds, see Option B. The resistors are calculated as follows:

Equation 21.

Equation 22.

Equation 23.

The lower OVLO threshold is calculated from:

Equation 24.

When the R1 to R3 resistor values are known, the threshold voltages and hysteresis are calculated from the following:

Equation 25.

Equation 26.

Equation 27.

Equation 28.

Equation 29.

Equation 30.

10.2.1.2.6.2 Option B

If all four thresholds must be accurately defined, the configuration in Figure 22 can be used.

Figure 22. Programming the Four Thresholds

The four resistor values are calculated as follows:

• Choose the upper and lower UVLO thresholds (V_UVH) and (V_UVL).
Equation 31.
Equation 32.
• Choose the upper and lower OVLO threshold (V_OVH) and (V_OVL).
Equation 33.
Equation 34.
• When the R1 to R4 resistor values are known, the threshold voltages and hysteresis are calculated from the following:

Equation 35.

Equation 36.

Equation 37.

Equation 38.

Equation 39.

10.2.1.2.6.3 Option C

The minimum UVLO level is obtained by connecting the UVLO/EN pin to VIN as shown in Figure 23. Q1 is switched on when the VIN voltage reaches the POR_EN threshold (≊8.6 V). The OVLO thresholds are set using R3, R4. Their values are calculated using the procedure in Option B.

Figure 23. UVLO = POR_EN

10.2.1.2.6.4 Option D

The OVLO function can be disabled by grounding the OVLO pin. The UVLO thresholds are set as described in Option B or Option C.

For this design example, option B was used and the following options were targeted: V_UVH = 38 V, V_UVL = 35 V, V_OVH = 65 V, and V_OVL = 63 V. The V_UVH and V_OVL were chosen to be 5% below or above the input voltage range of 40 to 60 V to allow for some tolerance in the thresholds of the part. R1, R2, R3, and R4 are computed using the following equations:

Equation 40.

Nearest available 1% resistors should be chosen. Set R1 = 150 kΩ, R2 = 11.5 kΩ, R3 = 95.3 kΩ, and R4 = 3.74 kΩ.

10.2.1.2.7 Power Good Pin

The Power Good indicator pin (PGD) is connected to the drain of an internal N-channel MOSFET capable of sustaining 80 V in the off-state and transients up to 100 V. An external pullup resistor is required at PGD to an appropriate voltage to indicate the status to downstream circuitry. The off-state voltage at the PGD pin can be higher or lower than the voltages at VIN and OUT. PGD is switched high when the voltage at the FB pin exceeds the PGD threshold voltage. Typically, the output voltage threshold is set with a resistor divider from output to feedback, although the monitored voltage need not be the output voltage. Any other voltage can be monitored as long as the voltage at the FB pin does not exceed its maximum rating. Referring to the Functional Block Diagram, when the voltage at the FB pin is below its threshold, the 20-µA current source at FB is disabled. As the output voltage increases, taking FB above its threshold, the current source is enabled, sourcing current out of the pin, raising the voltage at FB to provide threshold hysteresis. The PGD output is forced low when either the UVLO/EN pin is below its threshold or the OVLO pin is above its threshold. The status of the PGD pin can be read through the PMBus interface in either the STATUS_WORD (79h) or DIAGNOSTIC_WORD (E1h) registers.

When the voltage at the FB pin increases above its threshold, the internal pulldown acting on the PGD pin is disabled allowing PGD to rise to V_PGD through the pullup resistor, R_PG, as shown in Figure 25. The pullup voltage (V_PGD) can be as high as 80 V, and can be higher or lower than the voltages at VIN and OUT. VDD is a convenient choice for V_PGD as it allows interface to low voltage logic and avoids glitching on PGD during power-up. If a delay is required at PGD, suggested circuits are shown in Figure 26. In Figure 26(A), capacitor C_PG adds delay to the rising edge, but not to the falling edge. In Figure 26(B), the rising edge is delayed by R_PG1 + R_PG2 and C_PG, while the falling edge is delayed a lesser amount by R_PG2 and C_PG. Adding a diode across R_PG2 (Figure 26(C)) allows for equal delays at the two edges, or a short delay at the rising edge and a long delay at the falling edge.

Figure 24. Programming the PGD Threshold

Figure 25. Power Good Output

Figure 26. Adding Delay to the Power Good Output Pin

TI recommends to set the PG threshold 5% below the minimum input voltage to ensure that the PG is asserted under all input voltage conditions. For this example, PGDH of 38 V and PGDL of 35 V is targeted. R5 and R6 are computed using the following equations:

Equation 41.

Equation 42.

Nearest available 1% resistors should be chosen. Set R5 = 150 kΩ and R6 = 10.5 kΩ.

10.2.1.2.8 Input and Output Protection

Proper operation of the LM5066I hot swap circuit requires a voltage clamping element present on the supply side of the connector into which the hot swap circuit is plugged in. A TVS is ideal, as depicted in Figure 27. The TVS is necessary to absorb the voltage transient generated whenever the hot swap circuit shuts off the load current. This effect is the most severe during a hot-short when a large current is suddenly interrupted when the FET shutts off. The TVS should be chosen to have minimal leakage current at V_IN,MAX and to clamp the voltage to under 100V during hot-short events. For many high power applications 5.0SMDJ60A is a good choice.

If the load powered by the LM5066I hot swap circuit has inductive characteristics, a Schottky diode is required across the LM5066I’s output, along with some load capacitance. The capacitance and the diode are necessary to limit the negative excursion at the OUT pin when the load current is shut off.

Figure 27. Output Diode Required for Inductive Loads

10.2.1.2.9 Final Schematic and Component Values

Figure 19 shows the schematic used to implement the requirements described in the previous section. In addition, Table 52 provides the final component values that were used to meet the design requirements for a 48-V, 10-A hotswap design. The application curves in the next section are based on these component values.

Table 53. Design Parameters

10.2.2.2.1 Selecting the Sense Resistor and CL Setting

LM5066I can be used with a V_CL of 26 or 50 mV. In general using the 26-mV threshold results in a lower R_SNS and lower I²R losses. This option is selected for this design by connecting the CL pin directly to V_DD. TI recommends to target a current limit that is at least 10% above the maximum load current to account for the tolerance of the LM5066I current limit. Targeting a current limit of 22 A, the sense resistor can be computed as follows:

Equation 43.

For this application, a 1-mΩ resistor can be used for a 26-A current limit.

10.2.2.2.2 Selecting the Hotswap FETs

For this design, the PSMN4R8-100BSE was selected for its low R_DSON and superior SOA. After selecting the MOSFET, the maximum steady-state case temperature can be computed as follows:

Equation 44.

Note that the R_DSON is a strong function of junction temperature, which for most D2PACK MOSFETs are very close to the case temperature. A few iterations of the previous equations may be necessary to converge on the final R_DSON and T_C,MAX value. According to the PSMN4R8-100BSE data sheet, its R_DSON doubles at 110°C. Equation 45 uses this R_DSON value to compute the T_C,MAX. Note that the computed T_C,MAX is already above the absolute maximum of the FET.

Equation 45.

This suggests that two FETs should be used for the design. During steady-state operation, the MOSFETs are fully enhanced and share current evenly. Thus, assuming that each FET carries half of the current, the T_C,MAX can be computed with Equation 46. Now T_C,MAX is 114°C, which is reasonable.

Equation 46.

10.2.2.2.3 Select Power Limit

In general, a lower power limit setting is preferred to reduce the stress on the MOSFET. However, when the LM5066I is set to a very-low power limit setting, it has to regulate the FET current and hence the voltage across the sense resistor (V_SNS) to a very-low value. V_SNS can be computed as shown in Equation 47.

Equation 47.

To avoid significant degradation of the power limiting, TI does not recommend a V_SNS below 4 mV. Based on this requirement, the minimum allowed power limit can be computed as follows:

Equation 48.

In most applications, the power limit can be set to P_LIM,MIN as shown. Here R_SNS and R_PWR are in ohms and P_LIM is in watts.

Equation 49.

The closest available resistor should be selected. In this case a 28.2-kΩ resistor was chosen.

10.2.2.2.4 Set Fault Timer

The maximum start time can be computed to 3.37 ms as shown in Equation 50.

Equation 50.

Note this start-time is based on typical current limit and power limit values. To ensure that the timer never times out during start-up, TI recommends to set the fault time (T_FLT) to be 1.5× t_start,max or 5.1 ms. This accounts for the variation in power limit, timer current, and timer capacitance. Thus, C_TIMER can be computed as follows:

Equation 51.

The next largest available C_TIMER is chosen as 100 nF. After C_TIMER is chosen, the actual programmed fault time can be computed as follows:

Equation 52.

10.2.2.2.5 Check MOSFET SOA

After the power limit and fault timer are chosen, it is critical to check that the FET stays within its SOA during all test conditions. During a hot-short, the circuit breaker trips and the LM5066I restarts into power limit until the timer runs out. In the worst case, the MOSFET’s V_DS equals V_IN,MAX, I_DS equals P_LIM / V_IN,MAX, and the stress event lasts for t_flt. For this design example, the MOSFET has 60 V, 4 A across it for 5.2 ms.

When the hotswap is in power limit and the FETs are operating in saturation region (V_GS close to threshold voltage), the designer cannot assume that the FETs will share. Even a small V_T mismatch causes a large difference in the current carried by the two MOSFETs. Thus, the SOA checking should be done assuming that all of the current is flowing through a single FET.

Based on the SOA of the PSMN4R8-100BSE, it can handle 60 V, 30 A for 1 ms and it can handle 60 V, 6 A for 10 ms. The SOA for 5.2 ms can be extrapolated by approximating SOA versus time as a power function as shown in the following equations:

Equation 53.

Note that the SOA of a MOSFET is specified at a case temperature of 25°C, while the case temperature can be much hotter during a hot-short. The SOA should be de-rated based on T_C,MAX using Equation 54.

Equation 54.

Based on this calculation, the MOSFET can handle 3.85 A, 60 V for 5.2 ms at elevated case temperature, but it is required to handle 4 A during a hot-short. In addition, there is tolerance on the power limit and timer, so using these settings would not produce a robust hotswap design.

10.2.2.2.6 Switching to dv/dt-Based Start-Up

For designs with large load currents and output capacitances, using a power-limit-based start-up can be impractical. Fundamentally, increasing load currents reduces the sense resistor, which increases the minimum power limit. Using a larger output capacitor results in a longer start-up time and requires a longer timer. Thus, a longer timer and a larger power limit setting is required, which places more stress on the MOSFET during a hot-short or a start into short. Eventually, there will be no FETs that can support such a requirement.

To avoid this problem, a dv/dt limiting capacitor (C_dv/dt) can be used to limit the slew rate of the gate and the output voltage. The inrush current can be set arbitrarily small by reducing the slew rate of V_OUT. In addition, the power limit is set to satisfy the minimum power limit requirement and to keep the timer from running during start-up (make P_LIM / V_INMAX > I_INR). Because the timer does not run during start-up, it can be made arbitrarily small to reduce the stress that the MOSFET experiences during a start into short or a hot-short.

The D2 prevents the charge of C_dv/dt from interfering with the power limit loop during a hot-short event and Q3 discharges C_dv/dt when the hotswap gate comes down.

10.2.2.2.7 Choosing the V_OUT Slew Rate

The inrush current should be kept low enough to keep the MOSFET within its SOA during start-up. Note that the total energy dissipated in the MOSFET during start-up is constant regardless of the inrush time. Thus, stretching it out over a longer time always reduces the stress on the MOSFET as long as the load is off during start-up.

When choosing a target slew rate, one should pick a reasonable number, check the SOA, and reduce the slew rate if necessary. Using 4 V/ms as a starting point, the inrush current can be computed as follows:

Equation 55.

Assuming a maximum input voltage of 60 V, it takes 15 ms to start-up. Note that the power dissipation of the FET starts at V_IN,MAX × I_INR and reduce to 0 as the V_DS of the MOSFET is reduced. Note that the SOA curves assume the same power dissipation for a given time. A conservative approach is to assume an equivalent power profile where P_FET = V_IN,MAX × I_INR for t = t_start-up / 2. In this instance, the SOA can be checked by looking at a 60-V, 1.76-A, 7.5-ms pulse. Using the same technique as section Check MOSFET SOA, the MOSFET SOA can be estimated as follows:

Equation 56.

This value has to also be derated for temperature. For this calculation, it is assumed that T_C can equal T_C,MAX when the board is plugged in. This would only occur if a hot board is unplugged, then plugged back in before it cools off. This is worst case and for many applications, the T_A,MAX can be used for this derating.

Equation 57.

This calculation shows that the MOSFET stays well-within its SOA during a start-up if the slew rate is 4 V/ms. Note that if the load is off during start-up, the total energy dissipated in the FET is constant regardless of the slew rate. Thus, a lower slew rate always places less stress on the FET. To ensure that the slew rate is at most 4 V/ms, the C_dv/dt should be chosen as follows:

Equation 58.

Next, the typical slew rate and start time can be computed to be 2 V/ms as shown in Equation 59, making the typical start time 30 ms.

Equation 59.

10.2.2.2.8 Select Power Limit and Fault Timer

When picking the power limit it needs to meet two requirements:

• Power limit is large enough to avoid operating with V_SNS < 4 mV
• Power limit is large enough to ensure that the timer does not run during start up. Picking a power limit such that it is 2× of I_INR,MAX × V_IN,MAX is good practice.

Thus, the minimum allowed power limit can be computed as follows:

Equation 60.

Next, the power limit is set to P_LIM,MIN using Equation 61. Here R_SNS and R_PWR are in ohms and P_LIM is in watts.

Equation 61.

The closest available resistor should be selected. In this case, a 28.2-kΩ resistor was chosen.

Next, a fault timer value should be selected. In general, the timer value should be decreased until there is enough margin between available SOA and the power pulse the FET experiences during a hot-short. For this design, a 10-nF C_TIMER was chosen corresponding to a 520 µs. The available SOA is extrapolated using the method described earlier.

Equation 62.

Next, the available SOA is derated for temperature:

Equation 63.

Note that only 4 A was required, while the FET can support 17.17 A. This confirms that the design is robust and has plenty of margin.

10.2.2.2.9 Chose Input and Output Protection and Set Undervoltage, Overvoltage, and Power Good Thresholds

This is identical to the previous design. Refer to Set UVLO and OVLO Thresholds, Power Good Pin, and Input and Output Protection for these settings.

10.2.2.2.10 Final Schematic and Component Values

Figure 40 shows the schematic used to implement the requirements described in the 48-V, 20-A PMBus Hotswap Design section. In addition, Table 54 provides the final component values used to meet the design requirements for a 48-V, 20-A hotswap design. The application curves in the next section are based on these components.