The TPS568215 operates using the adaptive on-time PWM control with a proprietary D-CAP3 control mode which enables low external component count with a fast load transient response while maintaining a good output voltage accuracy. At the beginning of each switching cycle the high side MOSFET is turned on for an on-time set by an internal one shot timer. This on-time is set based on the converter’s input voltage, output voltage and the pseudo-fixed frequency hence this type of control topology is called an adaptive on-time control. The one shot timer resets and turns on again after the feedback voltage (V_FB) falls below the internal reference voltage (V_REF). An internal ramp is generated which is fed to the FB pin to simulate the output voltage ripple. This enables the use of very low-ESR output capacitors such as multi-layered ceramic caps (MLCC). No external current sense network or loop compensation is required for DCAP3™ control topology.

The TPS568215 includes an error amplifier that makes the output voltage very accurate. This error amplifier is absent in other flavors of D-CAP3 control mode. For any control topology that is compensated internally, there is a range of the output filter it can support. The output filter used with the TPS568215 is a low pass L-C circuit. This L-C filter has double pole that is described in

Equation 1.

At low frequencies, the overall loop gain is set by the output set-point resistor divider network and the internal gain of the TPS568215. The low frequency L-C double pole has a 180 degree in phase. At the output filter frequency, the gain rolls off at a –40dB per decade rate and the phase drops rapidly. The internal ripple generation network introduces a high-frequency zero that reduces the gain roll off from –40dB to –20dB per decade and increases the phase to 90 degree one decade above the zero frequency. The internal ripple injection high frequency zero is changed according to the switching frequency selected as shown in table below. The inductor and capacitor selected for the output filter must be such that the double pole is located close enough to the high-frequency zero so that the phase boost provided by this high-frequency zero provides adequate phase margin for the stability requirement. The crossover frequency of the overall system must usually be targeted to be less than one-fifth of the switching frequency (F_SW).

Table 7-2 lists the inductor values and part numbers that are used to plot the efficiency curves in the Section 6.7 section.